диафрагмированные волноводные фильтры / 5ae274f5-adee-4a89-9a8e-78dfef57c204

Normalized Patterns (dB)

Dielectric Slab	Conductor Tape	Dielectric Slab

	Ferrite Slab		Ferrite Slab			Ferrite Slab
			(a)
0	0	30	H0 = 0 T		330	0
	330					30
			L	= 0.0 mm
	E-Plane		Lx	= 1.0 mm		H-Plane
-10	300		Lxx	= 1.5 mm
-20	270		90	270
-10	240		120	240
0	210	150			210	150
	180					180

(b)

Measured S11 (dB)

60

0

-5

-10 -15

-20

-25 (a)

-30 H0 = 0.0 T

5

-5

-10

-15

Lx = 1.5 mm

Lx = 1 mm Lx = 0 mm

90	-20	(b)		1,460 MHz
120	-25	H0 = 0.31 T
	-30
		9.0	9.5	10.0	10.5	11.0

Frequency (GHz)

(c)

Figure 4. (a) The fabricated ferrite-loaded bow tie slot antenna, (b) the normalized radiation pattern of the antenna at H0 = 0 T, and (c) the measured S11 of the antenna at bias magnetic fields of H0 = 0 and 0.31 T [53].

;S11; (dB)

		Radiating		Capacitor					H0 = 0 T,	C = 80 fF,		f = 10.82 GHz
		Slot							H0 = 0 T,	C = 67 fF,		f = 10.98 GHz
									H0 = 0.24 T, C = 67 fF,			f = 11.27 GHz
									H0 = 0.24 T, C = 50 fF,			f = 11.84 GHz
			SL							0	10
									30			-30
				Sw
											0
		CBCPW Via		Ferrite Slab					60		-10	-60
		Shielding
											-20
0		(a)							90			-90
										H-Plane
-5
										0	10
									30			-30
-10											0
							H0 = 0 T, C = 80 fF
-15							H0 = 0.36 T, C = 80 fF				-10	-60
-20							H0 = 0.32 T, C = 67 fF		60
							H0 = 0.36 T, C = 67 fF				-20
-25							H0 = 0.28 T, C = 50 fF
							H0 = 0.32 T, C = 50 fF
-30							H0 = 0.36 T, C = 50 fF		90			-90
10.5	11	11.5	12	12.5				13		E-Plane
		Frequency (GHz)

(b)

(c)

Figure 5. (a) The fabricated SIW antenna loaded with a capacitor and ferrite slab, (b) the measured S11 of the antenna with 2-D electrical and magnetic tuning, and (c) the measured radiation pattern of the antenna (dBi) [54].

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June 2015

metamaterial radiators to achieve a miniaturized CBS antenna. The zeroth-order resonance of the antenna is used for tuning, and a varactor diode is embedded within the IDC slot. By changing the applied voltage across the varactor diode, the capacitance value of the radiator changes slightly and a frequency tuning for the zeroth-order resonance in the range of 4.13–4.5 GHz (9%) is obtained. The gain of the antenna varies in the range of 2.5–4 dBi. The additional biasing slots on the top cavity wall might introduce additional leakage from the top side of the antenna. This becomes critical in terms of efficiency. However, the efficiency of the antenna is not studied in this article.

Ferrite-Loaded SIW Antennas (Based on Method IV)

Magnetically Tunable Ferrite-Loaded SIW Antenna

Tan et al. [52] proposed an SIW-CBS antenna loaded with ferrite slabs. Two ferrite slabs are inserted along the side walls of the SIW cavity slot antenna. Due to the changes in the magnetic biasing field, the operating frequency of the antenna is changed. A frequency tuning range of more than 10.5% at the X-band is achieved, and the antenna gain is more than 5 dBi within this range. Four neodymium external magnets are used to supply the bias magnetic field. The magnetic strength is tuned by changing the distance between the two magnets and the corresponding ferrite slab. The results presented in this article prove that the magnets do not affect the radiation characteristics of the antenna.

Ferrite-Loaded SIW Bow Tie Slot Antenna with Broadband Frequency Tunability

In [53], the method shown in Figure 1(d) and similar to what has been proposed in [52] is applied to an SIW cav- ity-backed bow tie slot antenna to make it tunable (Figure 4). Two ferrite slabs are inserted into the cavity and result in a tuning range of 14.95% in the X-band. The frequency of the antenna can be tuned in the range of 9.23–10.69 GHz with respect to changes in the bias mag-

While some tuning methods are based only on changing the electrical

characteristics of the cavity resonators, others aim to change both the magnetic and electrical characteristics.

netic fields in the range of 0–0.31 T. An antenna gain of more than 5 dBi over the tuning range is observed. Figure 4(a) shows the fabricated ferrite-loaded SIW bow tie slot antenna. The measured radiation pattern and S11 results are shown in Figure 4(b) and (c).

Simultaneous Electric and Magnetic

Two-Dimensionally Tuned ParameterAgile SIW Devices

The method shown in Figure 1(d) is used to achieve a simultaneous electric and magnetic tuning for an SIWCBS antenna in [54]. As a result, a wider tuning range is achieved in comparison with the case where only magnetic or electric tuning is used. A ferrite slab is inserted into the SIW cavity antenna, and then the frequency of the antenna is controlled by changing the external magnetic field. Using a method similar to the one proposed in [46], electrical tuning is also achieved. Instead of using a varactor diode, three different capacitor values are used as a proof of concept in the fabricated prototype. Applying both methods, a tuning range of 10.7–12.1 GHz is obtained. The fabricated prototype of the antenna is shown in Figure 5(a) along with the measured S11 and the radiation pattern of the antenna in Figure 5(b) and (c), respectively.

Tunable SIW VCOs

In this section, a review of the SIW-based VCOs reported in the literature is presented. Three different methods shown in Figure 1 are used to tune SIW resonators in oscillator circuits. Based on the methods used, a different tuning range, phase noise, dc power consumption, RF output power, and harmonic

Table 3. A detailed quantitative comparison for all of the mentioned VCOs.

									Second
Tuning				L at	L at			FOMT (dBc/Hz) at	Harmonic
Technique				100 kHz	1 MHz	Pdc	Pout	100 kHz/ FOMT	suppression
Based on	Reference	fc (GHz)	FTR	(dBc/Hz)	(dBc/Hz)	(mW)	(dBm)	(dBc/Hz) at 1 MHz	(dBc)
VCOs Method III	[54]	9.82	4.8%	-88	-117	37	6.5–10	165.59/174.6	35–50 dBc
	[55]	2.7, 3.7	N/A	-105.5	-119.5	160	5.33,	N/A	228.5
		dual band					10.83	FOM is +170
Method I	[56]	11.39	4.1%	-94	-124	20	1–2.9	174.37/177.37	—
Method V	[31]	12.2	2.45%	-98	-122	30	7–7.8	173.1/177.1	—
Method II	[57]	1.95	26%	-109	-130	51	4–5.1	186.42/187.42	228
June 2015									45

Vr

	(GHz)	10.0
		9.9
	Frequency	9.8
		9.7
	Oscillation	9.6
		9.5
		9.4
		9.30

					Vd
SIW Resonator		Cr	Cin
		Ground
		L1	S
			L3
	Varactor	G	D				Output
		L2	L4
			S
		Ground
		(a)
						10
						9	(dBm)
						8
							Power
		Measured Frequency
		Simulated Frequency				7
		Measured Output					Output
		Power
						6
5	10	15	20	25	305
	Reverse Voltage Vr (V)
		(c)

Phase Noise (dBc/Hz)

	(b)
100	60	(dBc)
95	50	Suppression
90	40
80	20 SecondHarmonic
85	30

9.3 9.4 9.5 9.6 9.7 9.8 9.9 Frequency (GHz)

(d)

Figure 6. (a) The physical configuration of the reflective VCO, (b) the fabricated VCO, (c) the measured and simulated oscillation frequency and the measured output power of the oscillator, and (d) the measured phase noise at a 100-kHz offset and the second harmonic suppression [55]. Vr: reverse voltage.

suppression is achieved for each design. Table 3 presents a detailed quantitative comparison for all of the mentioned VCOs.

Floating Patch Varactor-Loaded

Cavity (Based on Method III)

A Low-Phase-Noise VCO Using an Electronically Tunable SIW Resonator

In [55], the tuning method shown in Figure 1(c) is applied to an SIW cavity resonator. The tunable SIW cavity resonator is then used in a reflective type X-band oscillator and in series with a p-type high electron mobility transistor (HEMT) (Figure 6). Using the floating metal patch method, a tuning range of 4.8% along with a phase noise of around -88 dBc/Hz at a 100-kHz offset is presented. The dissipated dc power by the VCO is 37 mW. With an FOM of 184 dBc/Hz and an FOMT of 175 dBc/Hz at a 100-kHz offset, the SIW VCO shows potential for use in onboard applications. The layout of the VCO, a photo of the fabricated prototype, and the measured performance of the VCO are shown in Figure 6(a)–(d).

A Dual-Band Oscillator with Reconfigurable CavityBacked Complementary Split-Ring Resonator

Dong and Itoh [56] proposed an SIW dual-band oscillator based on a reconfigurable cavity-backed complimentary split-ring resonator. A method very similar to the one shown in Figure 1(c) and a p-i-n diode is used to make the SIW resonator reconfigurable. As a result, the oscillator has two different frequencies of operation. A switchable stub is used to make the matching possible for both states. A tuning ratio of 3.77:2.675 GHz is achieved. The phase noise at these two states is -99.6 and -105.5 dBc/Hz at a 100-kHz offset, respectively. The FOM of the SIW reconfigurable oscillator is +170 dBc/Hz at a 100-kHz offset.

Side Varactor-Coupled Cavity (Based on Method I)

Design of High-Q Tunable SIW Resonator and Its Application to Low-Phase-Noise VCO

The tuning technique shown in Figure 1(a) is applied to an SIW cavity resonator in [57]. The tunable SIW resonator is then used in a parallel feedback coupling oscillator design (Figure 7). The result is an SIW VCO with a tuning range of 4.1% (455 MHz at the X-band) with a phase

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June 2015

noise of -93 dBc/Hz at a 100-kHz offset. The dissipated dc power of the VCO is 20 mW. The FOMT of the VCO based on the reported data is 174.37 dBc/Hz at a 100-kHz offset. The measured results of the SIW resonator show an unloaded quality factor in the range of 286–299. The schematic and fabricated prototype photos of the SIW low-phase-noise VCO are shown in Figure 7(a) and (b), respectively.

Post-Loaded Mechanically Tuned Using Screw (Based on Method V)

Mechanically Tunable SIW Cavity-Based Oscillator

In [31], a different technique similar to what is shown in Figure 1(e) and first presented in [45] is used to tune a rectangular SIW cavity resonator. The method is based on a mechanically controlled flap and a metalized via post inside the cavity resonator. As the flap rotates around the axis of the via post, the connection

Using this method, the resonator can be tuned by 2–3%.

point of the via and the cavity changes, and, thus, there will be a shift in the resonance frequency of the resonator. Using this method, the resonator can be tuned by 2–3%. The designed resonator is then put in series with an HEMT to shape a reflective-type oscillator (Figure 8). The proposed VCO has a tuning range of 2.5% ( f0 = 12.4 GHz) and a phase noise of -122 dBc/Hz at an offset of 1 MHz. Dissipating 30 mW of dc power, the VCO has an FOM of 189.1 dBc/Hz at a 1-MHz offset. Figure 8(a) shows the fabricated prototype of the mechanically tuned SIW VCO. Measured and simulated operation frequencies of the VCO are shown in Figure 8(b). The measured phase noise of the VCO at different frequencies is shown in Figure 8(c).

12.3

2.5

(Unit: mm)

Vtune
MA46H120
2.5	0.4
			28 mm
0.3		3.0
12.3	4.5 2.6		Vdd
1.55
	MGF491AL		Vgg
2.5
	8.2	9
		2.
43 mm
(a)			(b)

;S11; (dB)

0
-5
-10
					Vtune = 0 V
-15					Vtune = 3 V
					Vtune = 6 V
					Vtune = 9 V
-20					Vtune = 12 V
				11.8	12.0	12.2
11.0	11.2	11.4	11.6

Frequency (GHz)

(c)

Output Spectrum (dBm)

0
-20
-60	Vtune = 0 V
	Vtune = 3 V
-80	Vtune = 6 V
-40	Vtune = 9 V

-100 Vtune = 12 V

11.111.2 11.3 11.4 11.5 11.6 11.7 11.8 11.9

Frequency (GHz)

(d)

Figure 7. (a) A schematic of the SIW-based VCO, (b) the fabricated prototype, (c) the measured S11 results of the SIW resonator, and

(d) the measured output spectrum of the SIW VCO [57].

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				Flap Bottom View
			(a)
		12.5
			Simulated
		12.4	Measured
(GHz)
		12.3
Frequency		12.2
		12.1
		12
			z = 0° z = 90°	z = 180°
			(b)
	-60			Measurements
	-70
(dBc/Hz)				Simulation
	-80			0°
	-90
Noise
	-100
Phase
	-110
	-120		180°
	-130
			105	106

Frequency Offset (Hz)

(c)

Figure 8. (a) The fabricated mechanically controlled SIW oscillator, (b) the simulated and measured output frequencies of the oscillator based on flap rotation, and (c) the SIW oscillator phase noise for different positions of the flap [31].

Via Post-Loaded SIW VCOs (Based on Method II)

A 1.7–2.2-GHz Compact Low-Phase-Noise VCO Using a Widely Tuned VCO Resonator

In [58], a new method was presented by which the authors targeted a wide tuning range for SIW-based VCOs. A widely tuned (+25%) SIW resonator is designed based on the technique described in Method I [Figure 1(b)]. However, instead of digitally tuning the resonator as in the cases of antennas and filters, a varactor diode is used to achieve analog tuning in this

Varactors’ dc

30 mm

Varactor Diodes	Tuning Vias
	Resonator
	Output
LPF	mm40
	pHEMT
	Transistor

VCO Output

Transistor’s dc

SubMiniature Version A

Top View Connector Bottom View

(a)

	2.3			Power		5.2
	2.2	Cv = 0.7 pF			(dBm)	4.8
(GHz)				Output
	2.1		Cv =			4.4
				1 pF
Frequency	2.0					4.0				2.1
	1.9	Cv = 1.6 pF				1.7	1.8	1.9	2.0
								Frequency
	1.8			Cv = 1.9 pF				(GHz)
	1.7						Cv = 2.4 pF
	1.6
		1	2	3			4		5	6

Cg (pF)

Phase Noise (dBc/Hz)

(b)

-107

-108

-109

-110

-111

1.71.8 1.9 2.0

Frequency (GHz)

(c)

	40	(dBc)
		Suppression
	32
	36
	28	Harmonic
	24	Second
	2.1

Figure 9. (a) The fabricated 1.7–2.2-GHz SIW tunable VCO, (b) the measured oscillation frequencies of the SIW VCO with respect to the changes in Cg and Cv (the inset shows the VCO output power for four different frequencies), and (c) the phase noise and second harmonic suppression for four different frequencies [58]. pHEMT: pseudomorphic high electron mobility transistor; LPF: low-pass filter.

case. The tunable SIW resonator is then employed in a reflective-type VCO. Due to the nature of reflectivetype VCOs, tuning ranges of more than 6% are difficult to achieve. While being tuned, the changes in the SIW

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June 2015

Omron Switch Top Via

dc Connector

resonator, used as a load for the VCO, will override the mandatory oscillation conditions. Similar to what has been proposed in [59], to overcome this problem, the authors employed another varactor diode at the gate of the HEMT. The task of this varactor is to compensate for the impedance changes of the SIW resonator. As a result of using the additional varactor diode in the circuit, the tuning range of the VCO is bounded just by the tuning range of the tunable SIW resonator. A tuning range of 1.7–2.2 GHz with a phase noise lower than -109 dBc/Hz at a 100-kHz offset is achieved. As a result of the unique tuning method used, the quality factor of the resonator and, consequently, the phase noise of the oscillator remain almost constant over the entire tuning range. Because of the two-layer structure used, the entire VCO circuit is placed on the backside of the SIW resonator. The proposed VCO has a phase noise of -109 dBc/Hz at a 100-kHz offset, and it dissipates 51 mW of dc power. The reported FOMT of the VCO is 186.42 dBc/ Hz at a 100-kHz offset.

Figure 9(a) shows the fabricated prototype of the VCO. The measured frequencies along with the output power are shown in Figure 9(b). The measured phase noise of the VCO along with the second harmonic suppression level is also shown in Figure 9(c).

The conventional tuning methods applicable to regular microwave structures are not effective for SIWs.

ing discussion. Table 4 presents a detailed quantitative comparison for all of the mentioned tunable filters. A complete review of SIW filters was recently presented in [49] where the focus was on studying and categorizing the tuning methods for SIW filters.

Via Post-Loaded

SIW Filters (Based on Method II)

A 1.2–1.6-GHz SIW RF-MEMS Tunable Filter

A two-pole, SIW, RF-MEMS cavity filter with 28% of tuning was presented in [41] (Figure 10). The tuning technique, as shown in Figure 1(b), is based on loading

Post D

Electrostatic Discharge

Protection Resistors (R1)

Post C

Reference A

Wall Via

Tunable SIW Filters

This section presents a review of the reported tunable SIW filters in the literature. Four of the tuning methods in Figure 1 have been used for SIW filters. Based on the different topologies used, different values for the tuning range, tuning frequency resolution, quality factor, and BW are obtained. While achieving a wide tuning range is very interesting, maintaining a high Q over the entire tuning range and/ or sometimes constant BW might be of more concern. A classification of the tunable SIW filters developed so far using these four methods is presented in the follow-

			mm				Bias Lines
			80
					114 mm
					SubMiniature Version A Connector
					(a)		Ref Al
	0				(T3)	0
	-10 (P1)
; (dB)	-20				; (dB)	-10
	-30
21					11	-20	(P1)			(T3)
;S	-40				;S
	-50					-30
	-601.0	1.2	1.4	1.6	1.8	1.0	1.2	1.4	1.6	1.8
		Frequency (GHz)					Frequency (GHz)
			(b)					(c)

Figure 10. (a) The fabricated tunable SIW filter with MEMS switches, (b) the measured S21, and (c) the measured S11 over the entire tuning range for the SIW filter [41].

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49

Table 4. A detailed quantitative comparison for all of the mentioned tunable filters.

Tuning
Technique
Based on	Reference	fc (GHz)	FTR	BW	Qu
Filters Method II	[40]	1.73	25%	2.3–3%	221–255
	[41]	1.4	28%	3.7%	93–132
Method III	[46]	2.76	8.69%	2.8%	—
	[60]	4.2	9.52%	1.5–2%	—
Method IV	[53]	+12.8	10%	Fixed BW of	+160
				+4.4% or
				tunable BW
				of 3–5%
	[61]	10.875	7.81%	—	—
		(simulation
		only)
Method VI	[47]	0.9	66%	4%	84–206

(a)

		0
		-5
	(dB)	-10
		-15
	Parameters-
		-20
		-25
	21	-30
	S
		-35
		-40

4 4.5 5 5.5 6 6.5 7 7.5 8 8.5 9

Frequency (GHz)

(b)

Figure 11. (a) The fabricated prototype of the varactor-loaded two-pole tunable filter and (b) the measured results at different states of 00, 01(10), and 11 [60].

each SIW cavity with perturbing via posts. A tuning range of 1.2–2.6 GHz is covered using 14 different tuning responses (states) with a very fine frequency resolution so as to behave such as a continuoustype filter. The insertion loss of the filter over this tuning range is below 4 dB. The magnitude of electric field distribution and resonance contour figures for the SIW cavity resonator are used to design a tunable filter with the highest tunability range possible. Packaged MEMS switches from Omron are used as the switching elements and are directly mounted on the biasing circuit layer of the filter. As a result, the parasitic effects are minimized. The quality factor of the filter varies from 93 to 132 over the tuning range. The stopband rejection performance is improved using two low-pass filters at the input/output ports of the filter. Figure 10(b) and (c) shows the measured S-parameter results for the presented RFMEMS tunable filter.

Tunable SIW Bandpass Filters with p-i-n Diodes

Armendariz et al. [40] presented a tunable SIW filter implemented using p-i-n diode switching elements. The perturbing via post method used in [41] is also used here. The biasing network and the SIW filter are completely separated using a two-layer structure. The performance characteristics of this filter are discussed in detail in [49] and are not repeated in this article.

Floating Patch Diode-Loaded Filters (Based on Method III)

Analog Tuning of Compact

Varactor-Loaded Combline Filters in SIWs

Sirci et al. [46] developed a two-pole tunable SIW coupled resonator filter. This filter is used as a demonstration of the tuning technique shown in Figure 1(c). Then, the application of this tuning method to the fine-tuning of the narrowband multipole filters in the postfabrication process is studied. The method enables fine-tuning of the center frequency without introducing degradation of the in-band performance. The filter in [60] also uses the same method. To initially evaluate the tuning concept, the authors fabricated a simple 2-bit tuned two-pole Chebyshev filter at

50

June 2015

Biasing Wire

Conductor Tape Varactor Diodes

l3	W	W	1

i

;Sij; (dB)

;Sij; (dB)

(1) l1

0

-10

-20

-30

-4011

0

-10

-20

-30

		Ferrite
W		Slab	Two 0.1-pF
	(2)	W2	Capacitors
			in Series
	l2	CBCPW to SIW
	(a)		(b)

				;S11;
				;S21;
				H0 = 0 T,		Vb = 0 V
				H0 = 0.24 T, Vb = 10 V
				H0 = 0.32 T, Vb = 20 V
11.5	12	12.5	13		13.5	14
		Frequency (GHz)
		(c)
				;S11;
				;S21;
				H0	= 0 T,	C = 50 fF
				H0	= 0.22 T, C = 75 fF
				H0	= 0.28 T, C = 80 fF

-4010 10.5 11 11.5 12 12.5 13 Frequency (GHz)

(d)

Figure 12. The fabricated SIW second-order Chebyshev bandpass filter with (a) a top view (varactor diodes) and (b) a bottom view (fixed capacitors), (c) the S-parameter results for the frequency-tunable filter case, and (d) the S-parameter results for the tunable BW filter case [54].

6 GHz [Figure 11(a)]. A tuning range of 5.5–6.2 GHz	Ferrite-Based SIW Tunable
with an insertion loss of lower than 3 dB for this fil-	Filters (Based on Method IV)
ter was measured. The measured results are shown in

Figure 11(b). The filter fabrication was realized using a single-layer SIW technology.

The filters in [46] and [60] were recently discussed in detail in a review paper on SIW filters [49]. Thus, this information is not repeated here.

Simultaneous Electric and Magnetic TwoDimensionally Tuned Parameter-Agile SIW Devices

The method of simultaneous electric and magnetic tuning of SIW structures is presented in [54] [shown in Figure 1(d)]. Magnetic ferrite slabs (YIG with 4rMs = 1,780 G

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51

wc = 1.53 cm wc = 1.53 cm	3 cm
Pd = 1.9 cm Pd = 1.9 cm

Qang = 100°	Qang = 100°
5 mm
	8 cm
	(a)
	Additional dc
	Bias Resistors

					(b)
	0
(dB)	-20
	-40
21
S	-60
	-80
	0
(dB)	-5
	-10
11	-15
S	-20
	-25	0.6	0.7	0.8	0.9	1.0	1.1	1.2	1.3	1.4
	0.5

Frequency (GHz)

(c)

Figure 13. (a) A designed and (b) fabricated octave tunable three-pole filter with measured (c) S21 and S11. [47]

and TH # 17 Oe) are used as the magnetic tuning element. A method very similar to that in [46] and [60] is also used to tune the SIW structures electrically. The concept of 2-D simultaneous tuning is applied to a two-pole passband filter. As a result of this combination of tuning methods, both frequency and BW agility is obtained. A tuning range of 10% can be obtained while maintaining a constant BW of 4.4% or a fixed frequency response with a tunable frequency BW of 3–5%. The insertion loss of the filter is lower than 4 dB. Two prototypes of the filter with both fixed capacitors and varactor diodes are fabricated. Figure 12(c) and (d) shows the measured S-parameter results for the filters in Figure 12(a) and (b) in the cases of frequency tuning and BW tuning, respectively.

Magnetically Tunable SIW Bandpass Filters Employing Ferrites

Almalkawi et al. [61] present an SIW double-circulator tunable filter in which a ferrite material is inserted into the waveguide. The method uses only the mag-

netic tuning approach similar to what has been shown in Figure 1(d). However, this article does not propose any measurement results. Thus, a tunability range of 10.45–11.3 GHz is achieved by changing the dc-mag- netic bias applied to the ferrite disk located at the edge of the SIW cavity based on simulations. The lowest internal bias (i.e., 2,100 G) required for saturating the ferrite magnet is applied to a single-circulator filter, and it is shown that by increasing the diameter of the ferrite disk for a given dc-magnetic bias, the BW of the passband filter decreases.

SIW Surface Ring-Gap Combline Filter (Based on Method VI)

Theory and Design of Octave Tunable Filters with Lumped Tuning Elements

The method shown in Figure 1(f) is applied to twoand three-pole SIW surface ring-gap combine filters to achieve a widely tuned bandpass filter [47]. This article presents the theory and design of octave tunable filters using this method in detail. The two-pole filter has a tuning range of 0.5–1.1 GHz and a measured insertion loss of 1.67 dB at 1.1 GHz. The three-pole filter has a tuning range of 0.58– 1.22GHzandameasuredinsertionlossof2.05dB.The3-dB fractionalBWofboththetwo-andthree-polefilteris +4%. Figure 13 shows the proposed three-pole filter and its measured S-parameter response over the tuning range of 0.6–1.2 GHz.

Miscellaneous SIW Structures

Due to their high quality factor and isolation from the surrounding environment, SIW tunable resonators are also a suitable choice for microwave devices other than filters, antennas, and VCOs. Some examples are phase shifters, isolators, and fluidically tuned structures. In this section, some of these devices are presented.

Ferrite-Based SIW Tunable Phase

Shifters and Isolators

Tunable Nonreciprocal Ferrite-Loaded SIW Phase Shifter

In [62], a ferrite slab is inserted near one of the side walls of an SIW where the magnetic fields are maximum. In response to the changes in the external magnetic field, the propagation constant inside the SIW is changed. This results in modifying the phase of the SIW section. The nonreciprocal phase shifter operates at the X-band and offers an insertion loss lower than 2 dB and a return loss higher than 10 dB. A phase shifting of more than 400° is achieved by the phase shifter at an existence of 0.24 T of externally applied magnetic field.

This article also presents an isolator as a possible application of the phase shifter. The isolator is designed based on the principle of a differential phase shift circulator. The phase shifter is loaded with two ferrite

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slabs along the side walls of an SIW. When biased in the opposite direction with 0.14 T, port 2 becomes isolated from port 1 and an isolation of 20 dB at 12.5 GHz is achieved.

Tuning Using Fluidics

Analysis of a Variable SIW Resonator Enabled by Dielectric Material Perturbations and Applications

Barrera and Huff [63] proposed a variable SIW resonator enabled by dielectric material perturbations. Fluidic dispersions, such as low dielectric oil and high dielectric particles, are used in a prototype SIW resonator. The prototype is used as a proof of concept for three different applications: 1) frequency tuning, 2) material measurements, and 3) fluidic sensing. The unique method used seems to be more useful for the two latter applications, but the study of these two applications is beyond the scope of this article; here the authors instead focus on the tuning application of this structure. This article also presents the circuit model and closed-form expressions for the resonant frequency and unloaded quality factor of SIW resonators loaded by a dielectric via post filled by different dielectric liquids.

In the first application, the theoretical values for the dielectric constant and loss tangent of the fluidic dispersion show a tuning range of 20% at the X-band. A via post is first drilled inside the cavity resonator, and then, by inserting different fluidic materials, the dielectric constant and loss tangent inside the via post are altered, and the frequency of the resonator changes in response to this material characteristic variation. The measurements show more losses for the material composing the fluidic dispersion, and thus, the Q drops down to ten. As a result, this tunable resonator is not practically suitable for designing tunable SIW filters or antennas. However, it is noted that employing better materials for the fluidic dispersions will greatly increase the performance as a tunable resonator.

Conclusion

Due to their high quality factor (hundreds), high power handling, and good isolation, SIW structures are excellent candidates for a variety of microwave devices, including filters, antennas, VCOs, and isolators. However, their use has been hindered due to their narrow BW and high sensitivity to the fabrication process. Tuning the SIW-based microwave structures has the advantages of covering more bands, the possibility of postfabrication fine-tuning, and less crosstalk sensitivity. However, the conventional tuning methods applicable to regular microwave structures are not effective for SIWs. Finding a way to tune these structures over a wide tuning range while maintaining their high Q seems to be very appealing and, at the same time, very

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challenging. To address all these points in more detail, the main focus of this article was on frequency-tunable/ reconfigurable SIW structures. The reported tuning methods applied to SIW structures so far are studied, and their pros and cons are discussed in detail. Also, separate libraries for SIW-based tunable filters, antennas, and VCOs are provided.

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