диафрагмированные волноводные фильтры / 520eb2e3-5e3a-49bd-b423-d073c90adfa2

Abstract—A filtering antenna with quasi-elliptic response based on SIW H-plane horn is presented. It is mainly composed of two symmetrical horn elements embedded with the trapezoidal dual-mode cavity. By exploiting the cross-couplings of dual-mode cavity, two radiation nulls are introduced on the gain curve, showing a quasi-elliptic filtering response without increase of horn size. Then, a modified transition is printed in front of the horn aperture on both sides of the substrate, which acquires wider impedance bandwidth and maintains low back-lobe radiation throughout the bandwidth. Besides, the proposed antenna is designed, fabricated, and measured at Ku-band. Two measured radiation nulls can be observed at 14.02 and 17.98 GHz, where the gain drops significantly to below -28 and -38 dBi, leading to high frequency selectivity. A wide measured impedance bandwidth of 12.2% and a large front-to-back ratio (FBR) of more than 20 dB are achieved. To the best knowledge of the author, it is the first time that the filtering horn antenna with radiation nulls, wide bandwidth and excellent radiation is designed.

Index Terms—Filtering antenna, H-plane horn, radiation nulls, substrate integrated waveguide (SIW).

I. INTRODUCTION

WITH the continuous development of wireless communication system, the antenna, as its key component, is required to be miniaturized, integrated and multifunctional. The filtering antenna is proposed as an effective solution, which integrates filtering performance into the antenna [1]. Some filtering antennas have been reported, such as filtering microstrip antennas array [2], dielectric resonator-based filtering antennas [3], substrate integrated waveguide (SIW) slot filtering antenna [4], and waveguide filtering antenna array [5]. Furthermore, radiation nulls [transmission zeros (TZs)] have been introduced to improve

frequency selectivity in [6]-[10].

Filtering horn antennas have also been presented. In [11], [12], the metallized posts are employed in the pyramidal horn antenna and the SIW H-plane horn antenna to achieve frequency selectivity function. A filtering antenna has been realized through SIW cavity frequency selective surface (FSS) covered at aperture of horn antenna [13]. In [14], the filtering horn antenna with band-stop characteristics has been achieved by inserting a split-ring resonator (SRR) into the metallic flare of the horn. In [15], to achieve filtering function, four rows of inductive metallized via holes are embedded into the flare

region of the horn antenna. These filtering horn antennas [11]-[15] realized frequency selectivity. However, these have a relatively poor frequency selectivity, and some of them [11], [13], [14] have a relatively large physical size. Although the size can be effectively reduced, the performance of SIW horn is deteriorated due to the severe mismatch and undesired back radiation. There have been many methods [16]-[19] for enhancing the matching and the radiation performances of conventional SIW horns. The bandwidth is from 14.1 to 16.6 GHz by loading triangular transition [18], but the front-to-back ratio (FBR) above 15 dB is only between 14.2 and 15.4 GHz.

In the design of the filter, it is well known that introducing TZs can improve the frequency selectivity and the out-of-band performance. As far as we know, the method of introducing radiation nulls in the filtering horn antenna has never been reported in the literature. This letter mainly aims to present a filtering SIW horn antenna with quasi-elliptic response, which provides better frequency selectivity than existing filtering horn antennas [11]-[15]. Meanwhile, the proposed design can provide the better performance of the matching and radiation. A trapezoidal dual-mode cavity is embedded into the flare region of the horn antenna, which introduces two radiation nulls on the gain curve. Then, a modified transition printed in front of the horn aperture realizes the matching within the broadband and the large FBR. A quasi-elliptic filtering SIW H-plane horn antenna with superior performance of matching and radiation is realized, which is of great significance for the miniaturization, integration and multifunction of radio frequency front-end.

II. DESIGN OF THE FILTERING ANTENNA

Manuscript received XXX; This work was supported in part by the National Key Research and Development Program of China under Grant 2020YFA0713900.

Mi-Mi Yang, Li Zhang, Yu Zhang, and Yong-Chang Jiao are with the National Key Laboratory of Antennas and Microwave Technology, Xidian

University, Xi’an, 710071, China (e-mails: mmyang12@126.com; lizhang@mail.xidian.edu.cn; zy_ruoqing@163.com; ychjiao@xidian.edu.cn).

Hong-Wei Yu is with the 38th Research Institute of China Electronic Technology Corporation, Hefei, 230031, China (e-mails:lanceaaa@126.com).

Fig. 1. Geometry of the proposed filtering antenna with w1=10.5, w2=10.5, w3=26.3, l1=12.3, l2=30.6, l3=14.6, l4=8.4, d1=0.8, t1=2.5, t2=6.2, t3=4.6, t4=5.7, a1=6.9, a2=5.4, a3=7.1, g1=0.2, w4=6, w5=3, w6=1.7, l5=10, s0=0.5, s1=0.1, s2=0.5, s3=0.6, s4=0.9, c1=c2=c3=c4=0.5. (Units: mm)

The geometry of proposed filtering horn antenna is shown in Fig. 1. It is mainly composed of two symmetrical horn elements embedded with the trapezoidal dual-mode cavity, which are connected through a filtering power divider cavity. Horn aperture element is used to radiate and as the last resonator. In addition, a modified transition is printed in front of the horn aperture on both sides of the substrate, which acquires wider bandwidth and maintains low back-lobe radiation throughout the bandwidth. The RT/Rogers 5880 substrate (εr = 2.2 and tan δ = 0.001) with a thickness of 1.524 mm is used. The metallized vias are designed with diameter d=0.5 mm and pitch size p=0.8 mm to prevent the undesirable leakage.

A. Quasi-elliptic Response Using Trapezoidal Dual-mode Cavity

One metallized via is set in the trapezoidal dual-mode SIW cavity to perturb the modes. Fig. 2 demonstrates the electric field distributions of the modes, which are TE101, TE102, and TE201, respectively. A trapezoid dual-mode single cavity filter fed by two 50-Ω microstrip lines and its corresponding coupling topology are shown in Fig. 3(a). Node 1 and 2 represent TE102 and TE201 respectively. Node N represents TE101. Node S and L represent source and load respectively. In [20], it has been indicated that there will be a TZ in the cure of |S21| when a nonreasoning node (NRN), resonating at lower than passband, is coupled with an in-band node. The coupling coefficients of NRN and in-band node must be all positive signs to place a TZ below the passband, while an opposite one need to be mixed for the TZ above the passband. In our design, both

TE102 and TE201 are selected as the in-band modes, while TE101 is acted as the NRN, which is lower than the in-band resonant

frequency. In Fig. 2, it can be seen that the signs of coupling coefficients for TE102 at the input/output ports are consistent, while those of TE201 are opposite. Therefore, the TZ below the passband can be provided by the cross-coupling between TE101 and TE102 modes, while the TZ above the passband is achieved through the cross-coupling between TE101 and TE201 modes.

Frequency response of trapezoidal single-cavity filter is shown

in Fig. 3(b). It is clearly shown that one TZ is below the TE102, and another one is above the TE201, which proves the quasi-elliptic response with TZ at each side of the passband can

be obtained by the trapezoidal dual-mode cavity.

B. Coupling topology analysis of quasi-elliptic filtering horn antenna

Design-1 composed of two symmetrical trapezoidal dual-mode SIW cavities connected through a filtering power divider cavity, using waveguide ports to excite, is shown in Fig. 4(a). The simulated |S21| results of Design-1 under weak couplings are shown in Fig. 4(b). When a1=w1, there are TE102 and TE201 of the trapezoidal dual-mode cavity are working in the passband. But when a1<w1, the third transmission pole appears, indicating that this filtering power divider cavity can be designed as a resonator. It is also clearly shown that two TZs locate at stop band, which further verifies the feasibility of using the trapezoidal double-mode cavity to realize the

(a) (b)

Fig. 4. Design-1. (a) Geometry. (b) |S21| under weak couplings.

(a) (b)

Fig. 6. Coupling topologies of (a) the proposed filtering horn antenna. (b) the corresponding fourth-order quasi-elliptic filtering antenna. (Node S: source, node 1: the filtering power divider, node 2/3/4: TE102/TE101/TE201 in trapezoidal dual-mode cavity, node 5: horn aperture element.)

quasi-elliptic response in array design. Further, this cavity can be completely embedded in the SIW horn antenna. Replacing Port 2 and Port 3 of Design-1 with horn radiators, Design-2 is shown in Fig. 5(a). The simulated |S11| result of Design-2 under weak couplings is shown in Fig. 5(b). It can be seen the fourth transmission pole appears, indicating that the horn is also a resonator. Therefore, the coupling topology of the proposed filtering antenna is shown in Fig. 6(a). Due to its symmetry, it can be equivalent to the topology in Fig. 6(b). Node 3 represents the NRN. The relation of coupling coefficients in Fig. 6(a) and Fig. 6(b) are shown in (1).

C. Improved SIW H-Plane Horn Antenna

As shown in Fig. 7(a), the Horn-1 is a simple SIW H-plane horn antenna. Inspired by [18] and [19], a modified transition printed in front of the horn aperture on both sides of the substrate is proposed for enhancing the performance of matching and the radiation in this letter. The transition consists of four groups of metal strips along the y-direction. The first full metallic strip is retained, mainly to achieve wider impedance bandwidth, while the other three groups are transformed into a tapered-ladder-like structure, mainly to enhance FBR. The improved SIW H-plane horn antenna is called Horn-2 and shown in Fig. 7(b). The effects of the modified transition on the reflection coefficient and the E- and H-plane radiation patterns are shown in Fig. 8. The SIW H-Plane horn antenna with the modified transition can realize wide bandwidth on 14.0-17.2 GHz. And the FBR is improved effectively. These results show that the modified transition is effective for the SIW H-Plane horn antenna.

(a) (b)

Fig. 7. SIW H-plane horn antenna. (a) Horn-1: without modified transition. (b) Horn-2: with modified transition.

SIW H-plane horn antenna is 14.0-17.2 GHz, which covers the 15.0-17.0 GHz. Therefore, it is possible to achieve a filtering horn antenna with passband of 15.0-17.0 GHz. The coupling matrix theory of the filter is still important to the filtering horn antenna, although their designs are not identical. In Section II-B, the coupling topology of the filtering horn antenna is analyzed. For a bandpass filter with passband of 15.0-17.0 GHz and two TZs at 14.0 and 18.0 GHz, a coupling matrix is shown in (2).

S and L represent source and load of filter, respectively. The diagonal elements Mii are calculated by

fi is the resonant frequency of the ith node [21]. The coupling coefficients and the external quality factor can be extracted from the following relation [22]:

filtering horn antenna can be obtained, and then the final size is optimized by HFSS. In order to obtain the same response as the filter, the radiation port of filtering antenna should have the same external quality factor of the filter. In our design, the optimized results of filtering antenna are basically consistent with the frequency response of the coupling matrix, shown in Fig. 9, which means the filtering antenna has an applicable loaded quality factor. This proves the feasibility of a quasi-elliptic filtering horn antenna by embedding a trapezoidal dual-mode cavity into the flare region of the horn.

(a) (b)

Fig. 8. Performance comparison of the Horn-1 and Horn-2. (a) Reflection coefficients. (b) Normalized radiation patterns at 16.0 GHz.

Fig. 9. Frequency response of filtering horn antenna and coupling matrix.

III. MEASURED RESULTS AND DISCUSSIONS

A prototype of the proposed filtering antenna is fabricated to validate the simulated results, as exhibited in Fig. 10. The measured |S11|, gain versus frequency and radiation efficiency are depicted in Fig. 11, which are in good agreement with the simulated results. The measured |S11| below -10 dB from 15.13 to 17.10 GHz is obtained. The measured gain varies from 10.6 to 13.3 dBi in the operating bandwidth, agreeing well with the simulated one. Two measured radiation nulls can be observed at 14.02 and 17.98 GHz, where the gain drops significantly to below -28 and -38 dBi, leading to high frequency selectivity. It can also be seen that the radiation efficiency is around 85% in the passband but it sharply drops to near zero at radiation nulls.

The radiation patterns of the E-plane and H-plane at 15.2, 16.0, and 16.8 GHz are illustrated in Fig. 12. It can be observed that stable end-fire radiation patterns across the bandwidth. The measured cross polarization level is less than -18 dB both in the E-plane and in the H-plane. In addition, FBR of larger than 20 dB is obtained. The proposed design is compared with other published horn antennas in Table I. The proposed design

provides higher frequency selectivity by introducing two radiation nulls into the gain curve without increase of horn size, which has never been reported in filtering horn antenna literature. It also provides a wider bandwidth and maintains a high FBR throughout the bandwidth.

IV. CONCLUSION

In this letter, a filtering antenna with quasi-elliptic response base on SIW H-plane horn is designed, fabricated, and measured. The quasi-elliptic response is achieved by the trapezoid dual-mode cavity embedded in the SIW horn without increasing the size of the horn. The proposed design provides good frequency selectivity while ensuring a wider bandwidth and a high FBR throughout the bandwidth, which is of great significance to the miniaturized, integrated and multifunctional development of wireless communication system.

Fig. 11. Measured |S11|, gain and efficiency of the proposed filtering antenna.

E-plane (c) H-plane

Fig. 12. Simulated and measured normalized radiation patterns of the proposed filtering antenna. (a) 15.2 GHz. (b) 16.0 GHz. (c) 16.8 GHz.