Vankka J. - Digital Synthesizers and Transmitters for Software Radio (2000)(en)

List of Abbreviations

ACI	Adjacent channel interference
ACLR	Adjacent channel leakage power ratio
ACP	First adjacent channel power
ADC	Analog-digital-converter
ALT1	Second adjacent channel power
ALT2	Third adjacent channel power
AM-AM	Amplitude-dependent amplitude distortion
AM-PM	Amplitude-dependent phase distortion
ASIC	Application specific integrated circuit
BiCMOS	Bipolar complementary metal-oxide-semiconductor
BPF	Bandpass filter
CALLUM	Combined analogue locked loop universal modula-
tor
CATV	Cable Television
CDMA	Code division multiple access
CF	Crest factor
CFBM	Cartesian feedback module
CIA	Carry increment adder
CIC	Cascaded-integrator-comb
CICC	Custom integrated circuits conference
CLK	Clock
CMOS	Complementary metal-oxide-semiconductor

xxiv	Abbreviations

CORDIC	Co-ordinate digital computer
CP	Carry Propagation
CS	Carry Save
CSD	Canonic signed digit
CSFR	Constant scale factor redundant
D/A	Digital to analog
DAC	Digital to analog converter
DAMPS	Digital-advanced mobile phone service
dB	Decibel
dBc	Decibels below carrier
dBFS	Decibels below full-scale
DCORDIC	Differential CORDIC
DCT	Discrete cosine transform
DDFS	Direct digital frequency synthesizer
DDS	Direct digital synthesizer
DECT	Digital enhanced cordless telecommunications
DEMUX	Demultiplexer
DFF	Delay-flip-flop
DFT	Discrete Fourier transform
DNL	Differential non-linearity
DPLL	Digital phase locked loop
DRC	Design rule check
DSP	Digital signal processing
EDGE	Enhanced data rates for global evolution
EER	Envelope elimination and restoration
EF	Error feedback
ETSI	European telecommunications standards institute
EVM	Error vector magnitude
FET	Field-effect transistors
FFT	Fast Fourier transform
FIR	Finite impulse response
FPGA	Field programmable gate array
GCD	Greatest common divisor
GMSK	Gaussian minimum shift keying

Abbreviations	xxv
GPRS	General packet radio service
GSM	Groupe spécial mobile
HPF	High-pass filter
HSCSD	High-speed circuit switched data
IC	Integrated circuit
IDFT	Inverse discrete Fourier transform
IEE	Institution of electrical engineers
IEEE	Institute of electrical and electronics engineers
IEICE	Institute of electronics, information and communica-
tion engineers
IF	Intermediate frequency
IIR	Infinite impulse response
IMD	Inter-modulation distortion
INL	Integral non-linearity
ISI	Inter-symbol interference
ISM	Industrial, scientific and medicine
ISSCC	International solid-state circuits conference
LE	Logic element
L-FF	Logic-flip-flop
LINC	Linear amplification with nonlinear components
LIST	Linear amplification employing sampling tech-
niques
LMS	Least-mean-square
LMS	Least mean squares algorithm
LO	Local oscillator
LPF	Low-pass filter
LSB	Least significant bit
LTI	Linear time invariant
LUT	Look-up table
LVS	Layout versus schematic
MAE	Maximum amplitude error
MSB	Most significant bit
MSD	Most significant digits
MUX	Multiplexer

xxvi	Abbreviations

NCO	Numerically controlled oscillator
NEG	Negator
NRZ	Non-return-to-zero
NTF	Noise transfer
OSC	Oscillator
P/I	Pipelining/interleaving
PA	Power amplifier
PAR	Peak to average ratio
PCDE	Peak code domain error
PFD	Phase/frequency detector
PLD	Programmable logic device
PLL	Phase-locked loop
PPM	Part per million
PSK	Phase shift keying
QAM	Quadrature amplitude modulation
QDDS	Quadrature direct digital synthesizer
QM	Quadrature modulator
QMC	Quadrature modulator compensator
QPSK	Quadrature phase-shift keying
R/P	Rectangular-to-polar
RF	Radio frequency
RLS	Recursive least squares algorithm
RLS	Recursive least squares
RMS	Root-mean-square
RNS	Residue number system
ROM	Read-only memory
RTL	Register transfer level
RZ	Return-to-zero
RZ2	Double RZ
RZ2c	Double complementary
SFDR	Spurious free dynamic range
SIR	Signal-to-interference ratio
SMS	Short message services
SNDR	Signal to noise and distortion ratio

Abbreviations

xxvii

SNR	Signal-to-noise ratio
TDD	Time division duplex
TDD-WCDMA	Time division duplex WCDMA
TDMA	Time division multiple access
TEKES	Technology development center
VCO	Voltage controlled oscillator
VHDL	Very high speed integrated circuit HDL
VHF	Very high frequency
VLSI	Very large scale integration
VMCD	Voltage Mode Class-D
WCDMA	Wideband code division multiple access
XOR	Exclusive or
∆¦	Delta sigma

Chapter 1

1. TRANSMITTERS

This chapter provides a basic introduction to transmitter architectures. The classic transmitter architecture is based upon linear power amplifiers and power combiners. Most recently, transmitters have been based upon a variety of different architectures including Envelope Elimination and Restoration (EER), polar loop, LInear amplification with Nonlinear Components (LINC), Combined Analogue Locked Loop Universal Modulator (CALLUM), LInear amplification employing Sampling Techniques (LIST) and transmitters based on bandpass sigma delta modulators.

1.1 Direct Conversion Transmitters

The principle of the direct conversion transmitter is presented in Figure 1-1 In direct conversion transmitters, the band limited baseband signals are converted directly up to the radio frequency with in-phase and quadrature carriers. The band-pass filter after the signal summation is used to suppress the

I(n)

D/A

90		PA

Q(n)

D/A

Figure 1-1 Direct conversion transmitter.

2

Chapter 1

out of band signals generated by the harmonic distortion of the carrier. The direct conversion transmitter is theoretically simple (no IF components) and potentially suitable for high integration level solutions. The drawbacks are: an I,Q mixer is needed at RF frequency, there is LO-leakage at RF frequency (filtering impossible) and VCO pulling.

The image rejection is given by

R		1		+ 2	∆ Gcos(∆θ ) + ∆G2		(1.1)
	10 log	(				),
	10	1		2	∆Gcos(∆θ ) + ∆G2

where ∆G is the gain mismatch and ∆θ is the phase mismatch. For instance, with a 5 degree phase mismatch and 0.1 dB amplitude mismatch, the maximum achievable single sideband suppression is only 27.2 dB, as shown in Figure 1-2.

The strong signal at the output of the power amplifier may couple to the local oscillator (LO), which is usually a voltage controlled oscillator, causing the phenomenon known as injection pulling [Raz98]. This means that the frequency of the local oscillator is pulled away from the desired value. The severity of the injection pulling is proportional to the difference between the frequency of the local oscillator and the frequencies at the output of the PA. By taking advantage of this, the problem of injection pulling can be alleviated by using an offset LO direct-conversion structure. In this structure, the carrier signal is formed by mixing two lower frequency signals. An additional band-pass filter is needed to filter away the undesired carrier at frequency. Another solution is to generate the LO signal from a lower frequency VCO by the frequency multiplication or from higher frequency VCO by frequency division. The VCO frequency is harmonically dependent on the

Phase Error [ ]

Image Rejection Ratio

12

10

							20 dBc
8
6
					25 dBc
4
			30 dBc
2			c
0 0
	0.2		0.4	0.6	0.8	1	1.2	1.4	1.6	1.8	2
					Amplitude Error (dB)

Figure 1-2 Image-Rejection ratio.

Transmitters

3

LO signal; the pulling rejection is not, therefore, as advantageous as in the offset VCO. Reported direct conversion transmitters using CMOS are, for example, [Ors99], [Lee01], [Ger01], and [Liu00]. The benefits and drawbacks of using direct conversion architecture in a transmitter are heavily dependent on the particular case, i.e. on application area, modulation method and technologies.

1.2 Dual-Conversion Transmitter

The injection pulling can also be avoided by using a dual conversion transmitter presented in Figure 1-3. In this structure, the baseband data is first upconverted to the intermediate frequency and then to the desired radio frequency. The dual conversion transmitter has advantages. First, the quadrature modulation is performed at the fixed lower frequency leading to the better matching between I and Q. Second, the additional attenuation of the adjacent channel spurs and noise may be achieved by using a band-pass filter at the IF. The hardware can be partly shared with the receiver (same oscillator frequencies). The drawbacks are complexity (more components), lower integration level, impedance matching required for external components and more power consumption. The stopband attenuation of the image reject filter at the RF frequency has hard requirements due to high frequency and the high attenuation factor because the signal component at the image frequency has the same power as the desired sideband. The first analog IF mixer stage of the transmitter in Figure 1-3 can be replaced with a digital quadrature modulator as shown in Figure 15-1.

1.3 Transmitters Based on VCO modulation

The constant envelope modulator can simply be implemented by direct modulation of a voltage controlled oscillator (VCO) [Bax99]. Ideally the VCO output frequency can be expressed as

Q(t)

90°

IF filter	image reject
	filter

I(t)

Figure 1-3 Dual conversion transmitter.

4	Chapter 1
fout fo + Kv Vtune	(1.2)

where f is the base output frequency of the VCO, Kv is the VCO sensitivity in Hz/V and Vtune is the input voltage that tunes the VCO. In principle, this produces a FM signal proportional to the modulating signal. There are many disadvantages, however, to this approach:

*Frequency drift: change in the VCO frequency due to tuning voltage

drift

*Frequency pushing: change in the VCO frequency due to change in the power supply voltage

*Load pulling: change in the VCO frequency due to change in the VCO

load

The change in VCO frequency can be compensated so that the receiving radio end tells the error to the transmitting radio, which tunes the modulating signal ( Vtune ) in order to compensate changes in the tuning slope. In wireless communication systems using time division multiple access (TDMA), such as DECT, data is transmitted in bursts with inactive periods in-between. Figure 1-4 presents a DECT architecture that utilizes these inactive periods between bursts to force the VCO frequency to match the desired channel frequency by a closed PLL [Bax99]. During transmit bursts the PLL loop is open and the incoming data modulates the VCO. Since the transmit burst duration is short (< 500 s) in DECT and the requirements on the frequency error are not very tight (<50 kHz) [Bax99], the frequency drift in the VCO during the burst can be made so low that it is tolerable. Frequency pushing caused by the switching and power ramping of the power amplifier (PA) is also a problem. Another more severe problem is frequency pulling caused by changes in the input impedance of the PA when it is switched or ramped. While these problems can be overcome in the DECT system, they render direct modulation unsuitable for standards that have strict frequency control

		data
	fr	LOOP	VCO	RF
		FILTER

n/n+1

channel

Figure 1-4 VCO modulator architecture.

Transmitters

5

specifications, such as GSM [Bax99].

In an indirect modulation scheme, the problems of VCO drift and instability are overcome by digitally modulating a synthesizer rather than directly modulating a VCO as in a simple direct modulator. In indirect modulation, the modulating signal is injected while the PLL is closed [Bax99], this makes it possible to constantly maintain accurate frequency control. An indirect modulator architecture is illustrated in Figure 1-5 [Ril94], [Per97], [Bax01], and [McM02]. The architecture comprises an FIR filter and a frequency synthesizer. The FIR filter filters the data bits. It consists of an oversampling counter, a ROM look-up-table and a small amount of random logic. The FIR filter taps stored in the ROM are quantised to single-bit. A reference frequency f is needed to phase-lock the VCO to a stable source. The delta-sigma modulator (∆Σ) and dual modulus divider comprise a frac- tional-N frequency synthesizer. The key feature of this synthesizer approach is that it uses a digital ∆Σ to generate a bit stream b(n), which embodies the higher resolution of the k-bit input within the long term average of b(n). By making the k-bit input to the ∆Σ a function of time, the instantaneous frequency can be directly manipulated.

The advantage of this technique is both that no mixers are needed to upconvert the modulating signal to the carrier frequency and that the RF signal is inherently band-limited to suppress noise. The disadvantage of this technique is that the modulation bandwidth must be less than the synthesizer bandwidth to avoid any loop suppression of the modulating signal. Since the synthesizer closed-loop bandwidth is usually narrow in order to suppress the quantization noise of the ∆Σ modulator, the maximum bandwidth is limited. This problem, however, can be tackled by equalizing the signal entering the

fr					LOOP	VCO	RF
					FILTER
Oversampling
Counter
					n/(n+1)
			2
LUT
ROM			k	∆Σ	1
				MOD	b(n)	Frequency
	z	-1	Delta Sigma			Synthesizer
FIR
				Modulator

Filter

-1

z	channel

data

Figure 1-5 Indirect GMSK modulator with ∆Σ−fractional-N-synthesizer.