Vankka J. - Digital Synthesizers and Transmitters for Software Radio (2000)(en)

Power Amplifier Linearization

47

[Wri92] A. S. Wright, and W. G. Durtler, "Experimental Performance of an Adaptive Digital Linearized Power Amplifier," IEEE Transactions on Vehicular Technology, Vol. 41, No. 4, pp. 395-400, Nov. 1992.

Chapter 3

3. DIGITAL COMPENSATION METHODS FOR ANALOG I/Q MODULATOR ERRORS

The block diagram of the I/Q modulator and correction network is shown in Figure 3-1. The I/Q modulator has phase imbalance, gain imbalance and DCoffset errors. With a careful layout design, the errors can be minimized, but they can never be completely nulled. The phase imbalance is caused mainly by the local oscillator and phase shifter, which does not produce exactly 90 degrees of phase shift between the two channels. The gain imbalance is caused mainly by the mixers, which are not exactly balanced. The sideband suppression is a function of both the gain and phase imbalance (see Figure 1- 2). The carrier suppression is a function of the DC offset between the inphase (I) signal and the quadrature (Q) signal. This offset can be compensated by altering the DC offset of the input signals.

The effects of the modulator errors on the constellation are shown in Figure 3-2, Figure 3-3 and Figure 3-4. The phase imbalance results in rotation of the axes in the I/Q coordinates as shown in Figure 3-2. The gain imbalance results in distortion of the signal and transforms the circular constellation in the I/Q coordinates to elliptical ones, as shown in Figure 3-3. The DC-offset shifts all sample points the same amount in the same direction, as shown in Figure 3-4. Together these imbalances cause the bit-error rate of the connection to increase. In [Cav93], a more detailed discussion about analog IQ modulator errors and their effects on the communications can be found.

The effects of the analog I/Q modulator errors are modelled using matrix notation. The quadrature modulator (QM) output of Figure 3-1, when the

nonidealities are taken into account, can be written as [Fau91]
		(t) MV		(t) Ma	(3.1)
where

50									Chapter 3
M		§α cos(φ / 2)		β sin(φ / 2) ·					(3.2)
		©α sin(φ / 2)		β cos(φ / 2)¹
		a =		§a1 ·			,		(3.3)
				©a2 ¹

where and β are gains of the I and Q channels, Vm(t) is the quadrature input, and φ is the phase split between the channels. a1 and a2 are the dc offsets of the channels. The time invariant signals are expressed as the length of two vectors composed of the in-phase and quadrature phase parts of the signals. The representation in (3.2) is called a symmetric form of the error; (3.4) is the same error presented in asymmetric form. The symmetric model [Cav93] differs from the symmetric model presented in [Fau91] in that the phase imbalance is attributed completely to the Q channel. In this case

M		§α	β sin(φ ) ·
		© 0	β cos(φ )¹	(3.4)

To compensate the errors, correction terms should be added so that errors presented in (3.1), (3.2) and (3.3) are nulled. The corrected output for the quadrature modulator compensator (QMC) output should be (if the QMC is in series with the QM)

vc (t) Cvd (t) + b Gĭvd (t) + b

(3.5)

where vd(t) is the compensator input, and

Correction Network

I(n) I'(n)

D/A

90

Q(n) Q'(n)

D/A

Gain

Controller A/D

Figure 3-1 Quadrature modulator and correction network.

Digital Compensation Methods for Analog I/Q Modulator Errors

51

In-Phase Component

2
1.5
1
0.5
0
-0.5
-1
-1.5
-2-2	-1.5	-1	-0.5	0	0.5	1	1.5	2
			Quadrature Phase Component

Figure 3-2 Phase imbalance causes rotation of axes in IQ plane. The dashed line represents original constellation.

In-Phase Component

2
1.5
1
0.5
0
-0.5
-1
-1.5
-2-2	-1.5	-1	-0.5	0	0.5	1	1.5	2
			Quadrature Phase Component

Figure 3-3 Gain imbalance transforms circular constellation into elliptical one.

In-Phase Component

2
1.5
1
0.5
0
-0.5
-1
-1.5
-2-2	-1.5	-1	-0.5	0	0.5	1	1.5	2
			Quadrature Phase Component

Figure 3-4 DC offset moves all points in one direction with the same amount.

52

Chapter 3

C Gĭ M 1

secφ

§

β cos(φ / 2)

β sin(φ / 2)·

α sin(φ / 2)

α cos(φ / 2) ¹

(3.6)

αβ

©

correspond to M in (3.2). The DC offset correction terms are

b

a

(3.7)

The gain imbalance is

G

cos(φ

/ 2)

§1 /α

0 ·

1 / β ¹

(3.8)

cos(φ )

© 0

and the phase imbalance is

ĭ

§

1

tan(φ / 2)·

©

tan(φ / 2)

1

¹

(3.9)

The compensator structure for the symmetrical model is presented in Figure

3-5. The cos(φ/2)/cos(φ) in (3.8) is ignored in Figure 3-5.

With (3.4) as a starting point, C becomes

C Gĭ M

1

secφ

§β cos(φ )

β sin(φ )·

α

(3.10)

αβ

©

0

¹

The gain imbalance is

G

§1 /α

0

·

(3.11)

©

/ β ¹

and the phase imbalance is

0

ĭ

§ 1

− tan(φ )·

(3.12)

©0

sec(φ ) ¹

The compensator structure for the asymmetrical model is presented in Figure

3-6.

In a wideband transmitter, the frequency dependence of I/Q mismatches must be taken into account. The wideband correction system is similar to the narrowband systems shown in Figure 3-5 and Figure 3-6 except that the correction terms are replaced by FIR filters, and extra delays are added due to the non-causality of the FIR filters [Pun00].

3.1 Quadrature Modulator Errors Compensation

The linearizer circuits such as mapping predistorter or Cartesian coordinate negative feedback linearizer can adapt for the quadrature modulator imperfections (see Section 2.3.2 or 2.2) [Cav90]. The linearizer types that assume the distortion to be dependent only on the magnitude of the signal and not its phase cannot be used for modulator error compensation (see Section 2.3.3).

There are analog and digital compensation methods for quadrature modulator errors. The majority of the articles suggest an adaptive way of compen-

Digital Compensation Methods for Analog I/Q Modulator Errors

53

sating the I/Q modulator errors. There are two main ways to do this. One way is to use a particular training signal and the DSP processes the sampled information of this signal to cancel the changing nonlinearities. This method is used in [Fau91], [Fau92], [Loh93], [Cav91], [Cav93], [Jui94], [Hir96], [Cav97], [Pie01], [McV02]. The second way is to sample the data and compensate the errors as the data is transmitted or received without a test tone, as in [Hil94], [Cav97], [Mar00], [Ger92], [McV02]. A simple analog method of compensating the DC offset is presented in [Ris01], [Man9] and [Aki00].

The basic idea of adaptation is to have some samples of the data, and some sort of estimation of the imbalances. A digital signal processor calculates the correction terms to compensate for the I/Q modulator errors. In the literature, there are some algorithms used commonly when searching convergence of the correction terms. The choice of the adaptation algorithm leads to different convergence speeds and computational complexities in general, as well as to different remaining steady-state errors [Hay91]. These algorithms can be divided into gradient and surface fit (LMS, RLS) algorithms. The Newton-Rhapson algorithm used in [Loh93] is a gradient based. This method is susceptible to quantization and modeling errors [Cav97]. The least mean square (LMS) algorithm used in [Cav93], [Mar00] is simple to implement and is not computationally demanding. It is widely used because of its simple and effective nature. However, the problem with the LMS is its slow convergence. The recursive least squares (RLS) algorithm used in [Hil94] is computationally more demanding than the LMS, but the RLS method reaches convergence faster than the LMS.

The methods in [Fau91], [Fau92], [Loh93], [Cav93], [Hil94], [Cav97], [Mar00] use a feedback network as shown in Figure 3-1. At the QM (or PA) output is an envelope detector that operates at radio frequency. The detector has diode characteristics. This diode, however, has some transfer characteristics that must be taken into account. The correction terms vary with temperature and applied carrier frequency often making readjustment necessary.

3.1.1 Symmetric Compensation Method

In [Fau91], an automatic method of compensating modulator errors is proposed. The principle of the method can be seen in Figure 3-1. The method is based on baseband preconditioning, meaning that adjusting the baseband drive signals feeding the mixers compensates the errors. The output signal is measured with a diode detector and converted to digital form. Then, using this digital information, DSP searches compensation terms that are inputted to the correction network. It is possible to keep all the corrections independent of each other by applying them in the correct order [Fau91]. In [Fau91],

54

Chapter 3

the DC offset is cancelled first. The differential gain errors should be compensated second, and the phase errors last.

The compensation method starts by initializing the correction circuit. Then test vectors are applied to the I and Q inputs, and the adjustments begin. First, carrier leak is corrected by zeroing the drive signals. The DC compensation terms in (3.7) are adjusted by, for example, a one-dimensional search. This happens by holding constant and adjusting 2 so that detector output is minimized. When local minimum is found, 2 is held constant and

is adjusted to find minimum. This is repeated until optimum is found [Fau91].

The gain error in (3.8) is compensated using the test vectors (I = A Q = 0 and I = 0, Q = A) applied to the channels, and measuring the outputs. The ratio of the output measurements is the gain mismatch. An iterative approach is used in [Fau91], thus avoiding the need for a calibrated detector. The gain of the channel with the larger output is decreased until the amplitudes of both channels are equal.

When channels are balanced, the phase error in (3.9) is compensated. The y stem is fed with a constant amplitude phasor (I = cos(2 ft), Q = sin(2 ft)). The cross-term tan(φ/2) of Figure 3-5 is adjusted iteratively so that the I and Q amplitudes become equal, and the output circular, as it is in an ideal case.

The 900 MHz quadrature modulator consisted of a 90º phase splitter, a combiner and two diode ring mixers fed at a +7dBm RF drive level [Fau91]. The uncorrected modulator had an overall differential phase error 8º, a differential gain of 0.5 dB, and carrier leak of -20dBm. After correction the above errors improved to 0.4º, 0.02 dB and -63dBm, respectively.

The algorithm in [Fau91] can be modified so that the test vectors are not needed. The trajectory of the incoming modulation is sampled. This is not,

1/α

i

-tan(φ/2)

-tan(φ/2)

q

1/β

i'

-a1

-a2

q

I/Q error correction	unwanted DC
	elimination

Signal flow graph of symmetrical IQ correction network.

Digital Compensation Methods for Analog I/Q Modulator Errors

55

however, tested in [Fau91].

A similar symmetric compensation method is used in [Loh93], but the Newton-Rhapson algorithm is used instead for acquiring the compensation terms. The controller stores the values inputted to the quadrature modulator and the values read from the envelope detector. Five sets of values are obtained for β (gains for both channels), a1 a2 (dc-offsets) and φ (phase imbalance between channels). Thus five equations are obtained for five unknown variables at different time instants. When the impairment values are known, both channels are predistorted and new values are measured; the impairment values are estimated as before. This compensation method, however, needs a perfect knowledge of the envelope detector parameters, thus limiting the usage of this method [Cav97].

In [Cav97], a symmetric model for both quadrature modulator (QM) and quadrature modulator compensator (QMC) are used. However, some improvements are made compared to previous methods. The channel imbalance and offset are measured simultaneously. The method reduces sensitivity to quantization noise in the feedback path, and thus coarse quantizers can be used.

3.1.2 Partial Correction of Mixer Nonlinearity in Quadrature

Modulators

A byproduct of the baseband correction techniques [Fau91] in the previous section is that the nonlinearity effects are made worse. The differential phase correction often increases the magnitude of the drive signals. The DC correction term moves the mixer bias point, which causes the positive part of the waveform to suffer more compression than the negative part, so the system

i i'

1/

-tan(φ) -a1

-a2

q q sec(φ)/β

I/Q error correction	unwanted DC
	elimination

Figure 3-6 Signal flow graph of asymmetrical IQ correction network [Cav93].

56

Chapter 3

is no longer balanced. This causes the growth of even order intermodulation products and the re-emergence of the carrier leak term with an amplitude that is dependent on the applied level of the signal.

In [Fau92], it was shown how the baseband correction of linear errors can be extended to include partial compensation of the third order nonlinearities in the mixer. An 8 dB improvement in mixer IM3 was obtained on a 900 MHz system.

3.1.3 Asymmetric Compensation Method

The method in [Cav93] is an improved version of the method presented in [Fau91]. The improvement suggests that a more economical asymmetric form of compensator that results in lower computational load and faster convergence should be used. The symmetrical correction network in Figure 3-5 requires 4 multipliers and 4 adders, and asymmetric in Figure 3-6 needs 3 multipliers and 3 adders [Cav93]. Moreover, a simplification is made and division with is neglected in (3.11), since it has no substantial effect on the results, it produces only scale change of a few percentage points. This therefore reduces the number of the multipliers to two. The adaptation algorithm itself has only to acquire four remaining imbalance compensation terms in (3.10). In [Cav93], the LMS algorithm is used, providing an effective way of achieving convergence.

The DC offset is eliminated by setting I = 0, Q = 0 and adjusting to minimize the envelope detector output. A detailed description of this can be found in [Cav93], but the principle is that the DC correction term is obtained in two steps. First, 4 measurements are made in the output with different values of , and a correction term is obtained from these measurements. The article suggests that the sum of squared errors between the measurements and the fitted surface of the input signal is minimized. Second, the measurements are made with the current optimal correction terms. The correction term is updated and the measurements are repeated until convergence is achieved. In practice, about eight measurements should be enough to obtain the terms.

The gain and phase imbalance compensation is achieved next. In asymmetric compensator gain and phase compensation, terms must be acquired simultaneously; [Cav93] provides an algorithm to do so. Four test signals are used with the same amplitude, but with four different phases. The adaptation is complete when all four phases give the same output from the envelope detector. The details are presented in [Cav93]. Fewer than eight iterations are needed to reach a steady state, and each iteration needs four power measurements at the envelope detector output.

Digital Compensation Methods for Analog I/Q Modulator Errors

57

The adaptation method in [Hil94] uses the same principles presented in [Cav93], but it concentrates on the adjacent channel power minimization. The QM and PA produces unwanted components on the radio frequencies and these components may affect the data transmission on adjacent channels. The method adjusts the correction coefficients in a way that the adjacent' channel power is minimized. However, the RLS algorithm used requires more memory and computational capacity to achieve good results, and thus modern digital signal processors must be used and if LUT based predistortion is used the method is far too slow [Cav97].

3.1.4 Digital Precompensation Method without Training Signal

The training signal is fed to the QM input and then read out from the QM (or PA) output, and changed again to digital form. The training signals are generated by the transmit modem and they are perfectly known. However, when using training signals, the compensation must be performed so that the training data is not transmitted. This can be achieved in, for example, the manufacturing/testing phase, when the transmitter is tested before being sold to the customers. Testing with training signals can safely be carried out when the transmitter (phone etc.) is turned on and a certain amount of time spent on the training signals before the device starts transmitting data. If the training signals are used in the middle of transmission, it must be ensured that the data is not transmitted.

[Cav97] suggests compensation methods with and without training signals. The problem with the technique without a training signal is that the loop-delay of the feedback loop is unknown. The modulator and the envelope detector, as well as the A/D converters are not infinitely fast and thus a delay is added for the signal. There are, however, compensation methods for loop-delay; [Cav97] suggests a way that does not need extra computation for the time delay parameters. Compensation from random data results in slower convergence than with training signals, but it can be accomplished in the middle of transmission without affecting the transmission. It can thus be used all the time and it can adapt to rapid changes in the temperature or other variables.

The precompensation method presented in [Mar00] does not need a training signal. The patent in [Xav01] is similar to this method. In [Mar00], for simplicity, DC, gain and phase imbalance are treated separately, but the adaptation procedure can be made concurrently. The gain of the modulator- amplifier-detector chain must be known exactly in order to achieve perfect adaptation in [Mar00]. [Mar00] uses the LMS algorithm that produces convergence in 2000-2500 iterations. However, the author accepts that the physical system is not as good as simulated, since some analog effects limit